Voltage regulation is commonly required to prevent variation in the supply voltage powering various microelectronic components such as digital ICs, semiconductor memories, display modules, hard disk drives, RF circuitry, microprocessors, digital signal processors and analog ICs, especially in battery-powered applications such as cell phones, notebook computers and consumer products.
Since the battery or DC input voltage of a product often must be stepped-up to a higher DC voltage, or stepped-down to a lower DC voltage, such converters are referred to as DC-to-DC converters. Step-down converters, commonly referred to as Buck converters, are used whenever a battery's voltage is greater than the desired load voltage. Step-down converters may comprise inductive switching converters, capacitive charge pumps, and linear converters. Conversely, step-up converters, commonly referred to boost converters, are used whenever a battery's voltage is lower than the voltage needed to power its load. Step-up converters may comprise inductive switching converters or capacitive charge pumps.
Another type of converter may operate as either a step-up or a step-down converter, depending on whether the power input to the converter has a voltage above or below its output voltage. Commonly referred to Buck-boost converters, such circuitry is needed whenever a converter's input and output voltages are similar, such that variations in the input voltage preclude the use of a simple boost or Buck converter.
One example an application requiring both step-up and step-down conversion is supplying a regulated 3.3V output from a lithium ion (Lilon) battery. Since a Lilon battery exhibits a terminal voltage which decays from 4.2V when fully charged to below 3V when discharged, the converter must be able to step-down initially and step-up later.
Inductive Switching Converters
Of the above voltage converters, the inductive switching converter can achieve superior performance over the widest range of currents, input voltages and output voltages. The principles of inductive switching converter operation are described in application Ser. No. 11/890,818, titled “High-Efficiency DC/DC Voltage Converter Including Down Inductive Switching Pre-Regulator And Capacitive Switching Post-Converter,” filed contemporaneously herewith and incorporated herein by reference. Two examples of non-isolated inductive switching converters, a synchronous Buck step-down converter and synchronous boost step-up converter, are shown in FIGS. 1A and 1B.
Synchronous Buck converter 1 of FIG. 1A comprises a power MOSFET 3, an inductor 5, a synchronous rectifier power MOSFET 4, with a rectifier diode 8, and a capacitor 6. Operation of MOSFET 3 is controlled by a pulse-width modulation (PWM) control circuit 2, driving the gate of MOSFET 3. The gate drive may vary in polarity and voltage depending on whether MOSFET 3 is N-channel or P-channel. Synchronous rectifier MOSFET 4, generally an N-channel MOSFET, is driven out of phase with MOSFET 3, but MOSFET 4 is not necessarily on the entire time when MOSFET 3 is off. In general, MOSFET 4 conducts only during times when diode 8 is conducting.
While the control circuit controlling the operation of converter 1 is referred to as PWM control, implying a fixed-frequency variable-pulse-width operation, it may alternatively operate in a variable frequency mode where the clock period is allowed to vary, or alternatively alternating between varying and fixed frequency modes depending on load and input conditions.
The energy input from the power source, battery or power input into DC/DC converter 1 is switched or gated through MOSFET 3. With its positive terminal connected to the battery or input, MOSFET 3 acts like a “high-side” switch controlling the current in inductor 5. Diode 7 is a P-N junction parasitic to MOSFET 3, in parallel to its drain and source, which remains reverse-biased in normal operation. Since diode 7 does not carry current in normal operation, it is illustrated by dotted lines.
By controlling the current in the inductor 5 by controlling the on-time of MOSFET 3, the energy stored in the magnetic field of inductor 5 can be adjusted dynamically to control the voltage on output filter capacitor 6. The output voltage Vout is fed back to the input of PWM control circuit 2, which controls the current IL in inductor 5 through the repeated switching of MOSFET 3. The electrical load connected to the output of converter 1 is not shown.
Driven out of phase with MOSFET 3, synchronous rectifier MOSFET 4 conducts some portion of the time when MOSFET 3 is off. With its positive terminal connected to the inductor, i.e. to the node where the intermediate voltage Vx is present, and its negative terminal connected to the circuit ground, MOSFET 4 acts like a “low-side” switch, shunting the current flowing through diode 8. Diode 8 is a P-N junction parasitic to synchronous rectifier MOSFET 4, in parallel to its drain and source. Diode 8 conducts substantial current only during intervals when both MOSFETs 3 and 4 are off.
Both MOSFETs 3 and 4 are off during every switching transition to prevent shorting of the input power source. The so-called break-before-make (BBM) operation prevents shoot-through conduction by guaranteeing that both MOSFETs 3 and 4 do not conduct simultaneously so as to short or “crow-bar” the input terminal of converter 1 to ground.
During this brief BBM interval, diode 8 must carry the load current IL flowing through inductor 5. Unwanted noise can occur during the transitions associated with BBM operation.
If we define the duty factor D of converter 1 as the percentage of the time that energy flows from the battery or power source into DC/DC converter 1, i.e., the time during which MOSFET 3 is on, then the output-to-input voltage ratio of Buck converter 1 is equal to its duty factor:
            V      out              V              i        ⁢                                  ⁢        n              =      D    ≡                  t        sw            T      
This relationship for a Buck or synchronous Buck converter is illustrated by curve 17 in FIG. 2A in graph 15. Notice the Buck converter cannot smoothly reach a zero or unity transfer characteristic without exhibiting some discontinuities 19 and 21 at the extremes of D. This phenomenon occurs due to switching delays in the power MOSFETS and the control and gate drive circuitry of converter 1.
As long as the Buck converter's power MOSFET is still switching, tsw is limited to some portion of the clock period T, e.g. 5%<D<95%, essentially due to turn-on and turn-off delay within the MOSFET switch and its control loop. For example at a 95% duty factor and a 3 MHz clock, the off time for the high-side MOSFET 3 is only 5% of the 333 nsec period, or just 16 nsec. This means that MOSFET 3 must turn off and back in only 16 nsec—too rapidly to regulate over a 95% output-to-input conversion ratio. The minimum off-time problem impacts both synchronous and non-synchronous Buck converters. The problem is, however, further exacerbated in a synchronous DC/DC converter since no time remains for the synchronous rectifier MOSFET to turn on and then off again and still exhibit BBM operation.
Referring again to graph 15 in FIG. 2A, above some maximum duty factor Dmax, there is not adequate time to maintain switching operation, and converter 1 must jump from Dmax to a 100% duty factor, as shown by discontinuity 21. Above Dmax, converter 1 turns on the high-side MOSFET 3 and leaves it on for the entire period T. The abrupt transition 21 causes a glitch in the output voltage of converter 1. Moreover, at a 100% duty factor, Vout=Vin, as shown by line 16, and all regulation is lost as long as the switching is halted.
Synchronous boost converter 10 shown in FIG. 1B includes a low-side power MOSFET 12, a battery connected inductor 13, an output capacitor 15, and a “floating” synchronous rectifier MOSFET 14 with a parallel rectifier diode 16. The gates of the MOSFETs 12 and 14 are driven by break-before-make circuitry (not shown) and controlled by a PWM controller 11 in response to voltage feedback VFB from the output of converter 10, which is present across filter capacitor 15. BBM operation is needed to prevent shorting out filter capacitor 15.
The synchronous rectifier MOSFET 14, which may be an N-channel or a P-channel MOSFET, is considered “floating” in the sense that neith its source nor its drain terminal is permanently connected to any supply rail, i.e. to ground or Vbatt. Diode 16 is a P-N diode intrinsic to synchronous rectifier MOSFET 14, regardless of whether synchronous rectifier MOSFET 14 is a P-channel or an N-channel device. A Schottky diode may be included in parallel with MOSFET 16 but with series inductance may not operate fast enough to divert current from forward-biased intrinsic diode 16. Diode 17 is a P-N junction diode intrinsic to N-channel low-side MOSFET 12 and remains reverse-biased under normal operation. Since diode 17 does not conduct under normal operation, it is shown as dotted lines.
If we again define the duty factor D as the time that energy flows from the battery or power source into DC/DC converter 10, i.e. the time during which low-side MOSFET switch is on and inductor 13 is being magnetized, then the output-to-input voltage ratio of boost converter 10 is equal to the inverse of 1 minus its duty factor, i.e.
            V      out              V              i        ⁢                                  ⁢        n              =            1              1        -        D              =          1              1        -                              t            sw                    /          T                    
This relationship for a boost or synchronous boost converter is illustrated by curve 18 in FIG. 2A also in graph 15. Notice that boost converter 10 cannot smoothly reach a unity transfer characteristic without exhibiting some discontinuity at the lower extreme of D. This phenomenon occurs due to switching delays in the power MOSFET 12 and its control and gate drive circuitry.
As long as power MOSFET 12 of boost converter 10 is still switching, tsw is limited to some portion of the clock period T, e.g. 5%<D<95%, essentially due to turn-on and turn-off delay within MOSFET 12 and its control loop. For example at a 5% duty factor and a 3 MHz clock frequency, the off time for MOSFET 12 is only 5% of the 333 nsec period, or just 16 nsec. This means that MOSFET 12 must turn on and back off in only 16 nsec—too rapidly to regulate below a 5% output-to-input conversion ratio. The minimum on time problem impacts both synchronous and non-synchronous boost converters.
Referring again to graph 15 in FIG. 2A, below some minimum duty factor Dmin, there is not adequate time to maintain switching operation and converter 10 must jump from Dmin to 0% duty factor, as shown by discontinuity 20. Below Dmin, converter 10 turns on the synchronous rectifier MOSFET 14 and leaves it on for the entire period T. The abrupt transition 20 causes a glitch in the output voltage of boost converter 10. Moreover, at a 100% duty factor, Vout=Vin, as shown by line 16, all regulation is lost as long as the switching is halted.
So in both synchronous Buck converter 1 and synchronous boost converter 10, operating near a unity transfer characteristic, i.e. when Vout≈Vin, shown by line 16, is problematic.
The efficiency η of a voltage converter can be given by
  η  =                    P        out                    P                  i          ⁢                                          ⁢          n                      =                            I          out                ·                  V          out                                      I                      i            ⁢                                                  ⁢            n                          ·                  V                      i            ⁢                                                  ⁢            n                              
An analysis of inductive switching converter efficiencies is provided in the above-referenced application Ser. No. 11/890,818.
Graph 25 of FIG. 2B illustrates examples of typical conversion efficiencies for synchronous Buck and synchronous boost converters as a function of the converter's voltage conversion ratio Vout/Vin. As shown, line 26 illustrates the unity conversion condition, where Vout=Vin. Conversion ratios less than unity, on the left side of line 26, represent step-down conversion. Efficiency curve 27 represents an example of a Buck converter performing step-down voltage conversion. Conversion ratios greater than unity, on the right side of line 26, represent step-up conversion. Efficiency curve 28 represents an example of a boost converter performing step-up voltage conversion.
In general, boost converters exhibit lower efficiencies than Buck converters for comparable load currents, as illustrated by curves 27 and 28, primarily due to the fact that boost converters exhibit higher peak currents than Buck converters. This problem is further accentuated for high Vout/Vin voltage conversion ratios, especially for output voltages approaching ten times the input voltage, as illustrated by the decline of curve 28 with increasing conversion ratios.
Furthermore, in graph 25, efficiency curve 27 for Buck converters is not shown for conversion ratios below 0.1 and above 0.9 and likewise efficiency curve 29 for boost converters is not shown for conversion ratios below 1.1 and above 10, because it requires operation at a duty factor of below 10% or above 90%, an operating condition difficult to achieve, especially at high switching frequencies.
Buck-Boost Switching Converter (Prior Art)
The problem of non-isolated DC/DC switching converter operation near unity transfer is especially difficult in applications when the input voltage may vary above or below the desired output voltage. Examples of this application include the output of noisy AC adapters or in circuitry which must operate as a battery back-up during emergency conditions when a main source of power has failed.
Another scenario where a unity conversion ratio is required occurs when a battery's operating voltage range extends above and below the desired output voltage.
For example, the discharge of a Lilon battery starts at 4.2V at full charge, initially decays rapidly to around 3.6V, then decays slowly from 3.6V to 3.4V, and finally drops quickly to its cutoff at or below 3V. In the event that a DC/DC converter is needed to produce a well-regulated 3.3V output during this entire discharge period, a sub-unity conversion ratio of (3.3V/4.2V), i.e. a ratio of 0.79, is needed at the outset, indicating that a Buck converter is required. At the battery's end-of-life, the conversion ratio exceeds unity, becoming 3.3V/3V, i.e. a conversion ratio of 1.1, and this requires a boost converter to provide the desired 3.3V output voltage. Such an application demanding both step-up and step-down conversion requires a Buck-boost, or up-down converter.
In the case where the user wants to avoid the complexities of up-down conversion, one possible approach is to use only a Buck converter and give up some battery life by cutting the battery off early, e.g. at 3.3V. In practice, however, when considering battery manufacturing variations and converter drop-out and duty factor limitations, too much battery life is sacrificed to rely on a Buck-only converter solution.
If up-down conversion cannot be avoided, one possible solution involves Buck-boost conversion. A Buck-boost converter can easily be derived from combining synchronous Buck and boost converters into a merged circuit. In the circuit diagram of FIG. 3A, for example, a Buck-boost converter 35 comprises a synchronous Buck converter, comprising a P-channel or N-channel MOSFET 36, an inductor 38A, an N-channel synchronous rectifier MOSFET 37, an intrinsic rectifier diode 39, and a capacitor 44, is used to power a synchronous boost converter, comprising a low-side N-channel MOSFET 40, an inductor 38B, a synchronous rectifier MOSFET 41, an intrinsic rectifier diode 42, and a filter capacitor 43. Cascade Buck-boost converter 35 first steps down and regulates the input voltage to an intermediate voltage lower than the desired output voltage, and then steps this intermediate voltage up to produce Vout.
Conversely, in FIG. 3B a synchronous boost-Buck converter 45 comprises a boost converter, comprising a low-side N-channel MOSFET 46, an inductor 47, an N-channel or P-channel synchronous rectifier MOSFET 48A, an intrinsic diode 49, and a capacitor 54, which is used to power a synchronous Buck converter, comprising a MOSFET 48B, an inductor 52, an N-channel synchronous rectifier MOSFET 50, an intrinsic rectifier diode 51, and a filter capacitor 53, the combined cascade boost-Buck converter collectively driving a load (not shown). In this approach, the input voltage is first stepped-up to an intermediate voltage higher than the desired output voltage, and then is stepped back down to produce Vout.
The overall efficiency of either Buck-boost converter 35 or boost-Buck converter 45 is given by the product of the boost converter's efficiency ηboost multiplied by the Buck converter's efficiency ηBuck, mathematically as ηcascade=ηBuck*ηboost. Even if both converters are 85% efficient, the combined cascade converter reaches only a roughly 70% overall efficiency, significantly lower than the efficiency of a Buck converter or a boost converter operated alone. The overall loss in either a Buck-boost or boost-Buck cascade converter is worse than the loss in either a synchronous Buck converter or a synchronous boost converter, because there are more transistors in series between the input and output terminals, and because all the transistors are switching all the time.
As shown, boost-Buck converter 45 of FIG. 3B includes series-connected MOSFETs 48A and 48B with intermediate capacitor 54. Since in steady-state operation, the current in series-connected MOSFETs must be equal, MOSFET 48B is redundant and can be eliminated without impacting circuit operation. Even if this is done, boost-Buck converter 45 requires two inductors 47 and 52, a characteristic highly undesirable from a user's point-of-view.
Similarly, Buck-boost converter 35 of FIG. 3A includes inductors 38A and 38B with intermediate capacitor 44. Since in steady state operation the current in inductors 38A and 38B is the same, inductor 38B is redundant and may be eliminated without changing the function of the circuit. In fact, capacitor 44 may also be eliminated without significantly altering the operation of Buck-boost converter 35.
The resulting simplified Buck-boost converter 55, illustrated in FIG. 3C, comprises a single-inductor 59; four MOSFETs 57, 56, 60, and 61; diodes 58 and 62 and filter capacitor 63. The PWM control circuitry and break-before-make and gate buffer circuits are not shown. Depending on its terminal conditions, such a converter can operate in three distinct modes, Buck, boost, and Buck-boost.
In FIG. 3D, equivalent circuit diagram 65 represents the operation of Buck-boost converter 55 as a Buck converter, where MOSFETs 57 and 56 are switched out-of-phase under PWM control while MOSFET 61 remains turned-on, represented by resistor 67, and MOSFET 60 is turned off, represented by open circuit 66. The overall power loss in Buck-booster converter 55 is greater than in a synchronous Buck converter because it now includes the conduction loss in MOSFET 61, i.e. power lost continuously in resistor 67. As a result of this increased power loss, Buck-boost converter 55 operating in its Buck mode has a lower efficiency than conventional Buck converter 1 shown in FIG. 1A.
In FIG. 3E, equivalent circuit diagram 70 represents the operation of Buck-boost converter 55 as a boost converter, where MOSFETs 60 and 61 are switched out-of-phase under PWM control while MOSFET 57 remains turned-on, represented by resistor 71, and MOSFET 56 is turned off, represented by open circuit 72. The overall power loss in Buck-boost converter is greater than in a synchronous boost converter because it now includes the conduction loss in MOSFET 57, i.e. power lost continuously in resistor 71. As a result of this increased power loss, Buck-boost converter 55 operating in its boost mode has a lower efficiency than conventional boost converter 10 shown in FIG. 1B.
The loss of efficiency using Buck-boost converter 55 is illustrated in FIG. 4 in the plot of efficiency η for various output-to-input voltage conversion ratios Vout/Vin. For convenience, the efficiency curves 27 and 28 from FIG. 2B for conventional Buck and boost converters are repeated as curves 81 and 82, respectively, in FIG. 4.
Curve 83 illustrates the efficiency of Buck-boost converter 55 operating in Buck-only mode, shown in equivalent circuit 65. Because of the series resistance associated with on-state MOSFET 61, the efficiency of Buck-boost converter 55 in the Buck only mode is lower than that of the simple Buck converter (curve 81). This loss of efficiency can range from a few percent to over 10%, depending on operating conditions. Curve 85 illustrates the efficiency of Buck-boost converter 55 operating in full Buck-boost mode where all four switches are switching constantly. In this mode Buck-boost converter 55 exhibits even greater losses and poorer efficiency than Buck-boost converter 55 operating in Buck mode (curve 83).
Curve 84 illustrates the efficiency of Buck-boost converter 55 operating in boost-only mode, shown in equivalent circuit 70. Because of the series resistance associated with on-state MOSFET 57, the efficiency of Buck-boost converter 55 in the boost-only mode is lower than that of a simple boost converter (curve 82). This loss of efficiency can range from a few percent to over 10%, depending on operating conditions. Curve 86 illustrates the efficiency of Buck-boost converter 55 operating in full Buck-boost mode, where all four MOSFETs are switching constantly. In this mode, Buck-boost converter 55 exhibits even greater losses and poorer efficiency than Buck-boost converter 55 operating in boost mode (curve 84).
Operating near unity conversion ratios, where the output voltage is slightly above or below its input voltage, i.e. where Vout≈Vin, Buck-boost converter 55 must operate in the Buck-boost mode, where all four MOSFETs are switching constantly. The resulting efficiency (curve 87) can be 10% to 20% lower than the efficiency of conventional Buck and boost converters (curves 81 and 82).
The efficiency penalty for a voltage converter to be able to operate over a wide range of voltage conversion ratios using the prior-art Buck-boost converter is substantial. Moreover, the converter must change its operating mode whenever operating near unity voltage conversion ratios.
Charge Pump Converters
An alternative to the switched-inductor converter is a charge pump, a voltage conversion circuit using only switches and capacitors to perform voltage translation through repeated charge redistribution, i.e. the continuous charging and discharging of a capacitor network driven by a clock or oscillator.
The advantage of a charge pump is that at specific voltage conversion ratios, it can exhibit extremely high conversion efficiencies approaching 100%. The disadvantage is that it can only efficiently generate voltages that are exact integer multiples of the number of flying capacitors used in its converter circuit. Voltages other than select multiples exhibit low efficiencies.
A common charge pump 90 is illustrated in FIG. 5A where a single capacitor 93 is employed as a “doubler”, i.e. to double the input voltage. Charge pump 90 comprises four MOSFETs, 92, 91, 94 and 95, configured in a manner similar to an H-bridge except that one terminal, the source of MOSFET 95 is connected to the output terminal and reservoir capacitor 96 rather than to ground.
Operation of charge pump 90 involves repeatedly charging and discharging flying capacitor 93. During the charging phase, diagonal MOSFETs 94 and 91 are closed, charging capacitor 93 to the voltage Vbatt while MOSFETs 92 and 95 remain open. Thereafter, in the charge transfer phase, MOSFETs 94 and 91 are opened, MOSFETs 92 and 95 are closed, and energy is transferred from the flying capacitor 93 to the output reservoir capacitor 96, pumping the output voltage VCP to a value twice the battery voltage or 2·Vbatt 
The purpose of the MOSFET switch network is essentially to place the flying capacitor in parallel with the battery during the charging phase and in series, i.e. stacked on top of the battery's positive terminal, during the charge transfer phase, as illustrated by equivalent circuit 100 in FIG. 5B. In FIG. 5B, a voltage source 101 represents the battery input and a capacitor 102 charged to Vbatt represents flying capacitor 93. By stacking the voltages across voltage source 101 and capacitor 102 atop one another, the output voltage of the charge pump is the sum of the voltages, hence doubling the voltage input. The cycle then repeats with another charging phase.
FIG. 5C illustrates a charge pump 110 utilizing two flying capacitors 114 and 115 and a network of seven MOSFETs 111, 112, 113, 116, 117, 118 and 119. The purpose of the MOSFET switching network is to charge capacitors 114 and 115 in series, thereby charging each capacitor to one-half the battery voltage, i.e. Vbatt/2. During the charging of capacitors 114 and 115, MOSFETs 111, 112 and 113 are on and MOSFETs 116, 117, 118 and 119 are off. After the charging phase, the charged capacitors 114 and 115 are connected in parallel, and connected to the positive terminal of the battery. This connection is accomplished by turning on MOSFETs 116, 117, 118 and 119 and turning off MOSFETs 111, 112 and 113. The resulting output voltage, shown in equivalent circuit 121 of FIG. 5D, is equal to Vbatt+Vbatt/2, or 1.5Vbatt, as illustrated by battery voltage source 124 and the parallel combination of capacitors 122 and 123 stacked atop one another. Because the output voltage is equal to 1.5 times the input voltage, this charge pump is sometimes referred to as a “fractional” charge pump.
Actually, many different charge pump topologies are possible, but most use only one or two flying capacitors. A single flying capacitor charge pump is capable of efficiently delivering power at an output voltage equal to twice its input voltage, or alternatively, if during the charge transfer phase the capacitor is connected to the negative terminal of the battery an output voltage that is a mirror-image negative voltage of the battery, i.e. −Vbatt. In the latter configuration the charge pump is also known as an inverter. The inverter case is illustrated in equivalent circuit 130 of FIG. 5E, where the battery, represented by a voltage source 131, is used to charge a capacitor 132, and then, during the charge transfer phase, the positive terminal of capacitor 132 is connected to ground, i.e. the negative terminal of battery 131. Two-capacitor, fractional charge pumps may also be used to produce an output voltage equal to one-half the input voltage, as shown in equivalent circuit 135 of FIG. 5F where each of capacitors 137 and 138 are initially charged to one-half of the voltage Bbatt provided by voltage source 136 are then referenced to the negative battery potential (ground) to provide a positive potential equal to +0.5Vbatt, as shown, or alternatively to provide a negative, inverted potential equal to −0.5Vbatt (not shown).
One problem with charge pump converters is they operate efficiently only at conversion ratios equal to integral multiples of the number of flying capacitors; in other words, they are not true voltage converters. Specifically, if a desired load voltage Vout is below the voltage VCP that the capacitor network produces, the converter cannot adapt. To obtain a voltage-differential between the charge pump's output voltage VCP and the output voltage of the converter Vout requires a resistor or current source to support the voltage mismatch, and the voltage across that lossy element results in lost power and reduced efficiency. An analysis of charge pump efficiencies is described in application Ser. No. 11/890,941, titled “High-Efficiency DC/DC Voltage Converter Including Capacitive Switching Pre-Converter And Up Inductive Switching Post-Regulator,” filed contemporaneously herewith and incorporated herein by reference.
The efficiency of single-mode charge pumps is illustrated in graph 150 of FIG. 6A for charge pumps having various multipliers, including a doubler (curve 151), an inverter (curve 152), and fractional charge pumps (curves 153, 154 and 155). Curve 156 represents a direct battery connection, identical to a linear converter's maximum theoretical efficiency, i.e. assuming no quiescent operating current. In each case, as the input to output ratio approaches an integer multiple of ±½Vbatt, the efficiency increases. The charge pump is not capable of delivering an output voltage above that voltage, and to obtain a higher output voltage a charge pump having a different voltage multiplier, i.e. a different operating mode, must be employed.
Each curve shown in graph 150 represents a specific charge pump circuit, e.g. including those shown previously in FIGS. 5A-5F. Unless a load operates at an exact half-volt integral multiple of the input voltage, however, the efficiency of the charge pump converter using one or two capacitors will suffer. This behavior is especially problematic for battery powered products where the battery voltage changes markedly as the cell discharges. In the case of Lilon batteries, for example, the voltage can decay more than 1V during discharge, representing a 25% change. Therefore, even if the peak efficiency may be high at one specific operating condition and battery voltage, the overall efficiency of the converter averaged over the battery discharge curve is poor. Weighted average efficiencies can be lower than 60% using a single-mode charge pump.
One way to improve the average efficiency of a charge pump voltage converter is to switch modes between 1X, 1.5X and 2X automatically within one circuit. This feature is particularly useful to supply a fixed voltage over a wide input range. The efficiency of a mode-changing charge pump is illustrated in FIG. 6B, where as the battery decays the tri-mode converter circuit switches from 1X-battery-direct mode having an efficiency shown by curve 163, to a 1.5X-fractional-mode with efficiency shown by curve 162, and again to 2X-doubler-mode having an efficiency shown by curve 161. By switching modes in this zigzag pattern, the efficiency of the charge pump converter is improved because the output is not pumped to an excessively high value compared to the load.
Unfortunately, conditions still exist where the efficiency suffers substantially. The mode transitions exhibit dramatic shifts in efficiency (curve 163) at a conversion ratio of one, and again for curve 162 at a 1.5X conversation ratio. The mode transitions may also result in sudden current and voltage discontinuities, or produce instability or noise. To determine what conversion ratio is required, graph 160 also includes curves 166, 165, and 164 relating the required input voltage range (right hand axis) and conversion ratios to produce an output voltage of 3V, 3.5V and 4V, respectively.
Specifically, the charge pump converter in 1.5X mode does not perform well at conditions slightly above a unity conversion ratio, unfortunately manifesting even lower efficiencies than the above-mentioned inductive Buck-boost converter.
Dropout in Prior Art converters
Whenever the input voltage and the output voltage of a voltage converter approach one another within the range of several hundred milli-volts, e.g. Vout=Vin±200 mV, the quality of the converter's regulating ability suffers. Loss of regulation quality may be manifest in several ways, either by a one-time or repeated glitch or discontinuity in output voltage, by increased ripple, or by complete loss of regulation within some narrow voltage band. The phenomenon of degraded regulation whenever Vout approaches Vin is referred to as “dropout”, meaning the converter drops out of regulation.
The Buck converter of FIG. 1A and the boost converter of FIG. 1B both momentarily lose regulation as their switching duty factor jumps from Dmax or Dmin to 100% and they lose regulation completely while D=100%, since the input is essentially resistively connected to the output during the dropout condition.
While a Buck-boost converter does not exhibit permanent dropout, it can easily suffer a voltage glitch during mode transitions, whenever the converter switches its Buck mode to its Buck-boost mode, or from its Buck-boost mode to its boost mode. Mode transitions occur whenever the converter changes from a circuit having two power devices switching into one where four devices are switching, or vice versa.
To avoid the mode switching transition problem, a Buck-boost converter can be run continuously in Buck-boost mode with all four power devices switching continuously, but as shown in FIG. 4, its efficiency is then degraded under all input-output conditions and conversion ratios.
As stated above, a charge pump is incapable of regulating voltage without the use of a series-connected linear converter to provide the regulation function. Unfortunately, it is a well known phenomenon that all linear converters exhibit loss of regulation, i.e. dropout, whenever ΔV across their input and output terminals becomes too small. In essence, dropout occurs in a linear converter because the loop gain of the amplifier performing regulation drops precipitously as its transistor pass element changes from acting as a current source to acting as a variable resistor. If the pass element is a bipolar transistor, the loss of gain occurs at small values of VCE as the device transitions from its active operating region into saturation. In many bipolar linear converters, this dropout condition occurs at more than 400 mV.
In so-called “low dropout” linear converters or “LDOs”, a MOSFET capable of operating as a current source at a lower ΔV is substituted for the bipolar pass element, but the linear converter still drops out at 200 to 300 mV as the power MOSFET pass element transitions from its saturation, i.e. constant current, region into its linear, i.e. resistive, region of operation.
In conclusion, prior-art non-isolated high-efficiency converters exhibit dropout at voltage conversion ratios approaching unity. Mode switching, loss of regulation and dropout can be avoided only by sacrificing efficiency. Isolated converters such as flyback and forward converters are able to operate at high efficiencies near unity conversion without the need switching modes, but their use of physically-large tapped inductors, coupled inductors, and transformers precludes their application in most portable products.
Summary of Prior-Art Down-Up Converters
In conclusion, no existing charge pump converter, Buck-boost switching converter or other inductive switching converter is able to both step-up and step-down DC voltages efficiently, especially for conversion ratios near unity where Vin≈Vout. What is needed is an up-down converter that is efficient over a wide range of input and output voltages, and that does not need to change its operating mode as it operates near a unity voltage conversion ratio, i.e. when Vout≈Vin. Furthermore, the converter should be free from dropout problems, maintaining high-quality regulation even while biased with an output voltage within 200 mV of its input, i.e. within the range Vout=Vin±200 mV.